09 May 95 This power amplifier design offers not only extremely low distortion, but also operation in either Class-A or Class-B. In the Class-A mode, some new insights have been exploited to allow unusually linear handling of the situation if the push-pull current capability is exceeded. The theoretical basis of the design is explained in detail in the Electronics World article "A TRIMODAL Power Amplifier" published in two parts in May and June 1995. As before, the basic philosophy is the linearisation of each stage with a high degree of local negative feedback before closing the global feedback loop; this allows the global feedback factor to be kept relatively low to maximise HF stability, while still showing exceptionally good linearity. The main innovative features are the mode-switching capability, and the unusually low distortion in Class-AB. The PCB follows exactly the design published in Electronics World for May and June 1995, including the SOAR (safe operating area) protection against output short-circuits, and the inclusion of rail fuses. The SOAR protection locus allows output power to be increased up to approx 40W in 8-Ohms without premature protection coming in; however be aware that running the design in Class A/AB mode at this sort of power level is going to need very large heatsinks for the output devices. Please note that it is not in general possible for us to give advice on modifications to the circuitry. The board is of high-quality roller-tinned fibreglass format with a full silk-screen component ident and solder-mask to minimise the possibility of solder shorts. Improvements over the previous Class-B PCB have been made as follows: The form-factor of the PCB has been altered so that two will fit side by side in a 19" rack chassis. All transistor positions have emitter, base and collector marked on the top-print to aid fault-finding. TO3 devices are also identified on the copper side. Wire links have been numbered to make it easier to check they have all been fitted. CIRCUIT NOTES. The space available for articles in EW is a finite resource, and so extra information for which no room could be found is given here about the details of the circuitry and its operation. 1 THE INPUT STAGE. The input stage follows my design methodology in running at a high tail current to maximise transconductance, and then linearising it by adding input degeneration resistors R2,3 that reduce the final transconductance to a suitable level. A current- mirror TR10,11 forces the collector currents of the two devices to be equal, thus balancing the input pair and ensuring that it does not generate second-harmonic distortion; even a small Ic imbalance generates second-harmonic at a much higher level than the unavoidable third harmonic. The mirror is degenerated by R6,7 to minimise the effects of Vbe mismatches in TR10,11; the value of R6,7 is not critical and any value in the range 10 - 68 Ohms should be sufficient. As a result of the insights gained while studying the slew-rate characteristics of this configuration, I have increased the input- stage tail current from 4 to 6 mA, and increased the VAS current from 6 to 10 mA over the original Class-B circuit. This increases the maximum slew rate, though as described elsewhere  the full theoretical speed increase is virtually impossible to obtain. One reason is feedthrough in the VAS current source; in the original circuit an unexpected slew-rate limit was set by fast edges coupling through the current source c-b capacitance to reduce the bias voltage at the worst possible time. This effect is minimised in this design by using the negative- feedback type of current source bias generator, with VAS collector current as the controlled parameter. TR21 senses the voltage across R13, and if it attempts to exceed the Vbe, it turns on further to pull up the bases of TR1 and TR5. C11 filters the rail ripple from the supply to this circuit and prevents ripple injection from the V+ rail. The simple act of making the controlled quantity the VAS current rather than the input tail-current actually contains a beautifully subtle trap for the unwary. Distortion may be increased in one of two ways; either "molehill distortion" in which lumps appear on the distortion residual as the VAS approaches positive clipping, or a general introduction of second-harmonic. Which effect appears seems to vary from amplifier example to example and probably depends on beta spreads. Both effects are due to TR21 altering its collector voltage to keep the VAS current constant; the VAS operating conditions vary widely compared with the devices in the input stage, and the latter has its linearity degraded by the varying bias voltage. R5, C14 provide decoupling to remove these bias perturbations and give a complete cure to either manifestation of the problem. This effect seems too obscure and specialised to add to the list of Great Distortions, which has now reached eight. Note that the input pair positions on the PCB are laid out for base-in-the-middle TO92 transistor packages. If the high-beta 2SA970 or similar devices are used, they will be found to have base and collector swopped, so the transistor legs will need to be sleeved then crossed over. There are positions on the PCB for the input bootstrap components, (C15,R47) but these may simply be omitted if the facility is not required; for example, a competent preamp with a buffered output should be able to drive the 2K2 input impedance without problems. If desired R1,R8,R40 and C2 may be returned to the original values used in the Class-B amplifier, which gives a high input impedance without bootstrapping; this will of course degrade the DC offset and noise performance back to their original (but still very respectable) levels. 2 THE VOLTAGE-AMPLIFIER STAGE. (The VAS) The VAS is linearised by beta-enhancing stage TR12, which increases the amount of local NFB through Miller dominant-pole capacitor C3. (often referred to as Cdom in these writings) Note that R36 has been increased to 2K2 to minimise power dissipation, as there is no significant effect on linearity or slewing. Do not, however, attempt to omit it altogether, or linearity will be affected and slewing much compromised. I have tried replacing it with a constant-current source, but this seems to give no benefits in linearity. As described in , the simplest way to prevent ripple from entering the VAS via the V- rail is old-fashioned RC decoupling, with a small R and a big C. We have some 200 mV in hand in the negative direction, compared with the positive, and expending this on voltage-drop through the RC decoupling will give symmetrical clipping. R37 and C12 perform this function; the low rail voltages in this design allow the 1000uF C12 to be a fairly compact component. 470uF is in general adequate, but the slight increase in output ripple is measurable. (if not audible) 3 THE OUTPUT STAGE. The output stage is of the Complementary Feedback Pair (CFP) type, which gives better linearity and much better quiescent stability than the more common EF alternative, as a result of the local negative feedback loops around driver and output device. Where there is negative feedback there is always the possibility of oscillation, so this configuration is slightly less straightforward to apply, which may account for its lack of popularity. When oscillation in a CFP stage does occur, it will probably be at a much higher frequency than global-feedback Nyquist oscillation (say 2 MHz rather than 400 KHz, but these are VERY rough figures) and is often very sharply confined to one half-cycle only, due to the parameter differences that exist even between nominally complementary devices. I have always found that the 100-Ohm base- stopper resistors (R12,R24 in the circuit) are a complete preventative against this sort of misbehaviour, but it is possible that with some combinations of devices this value may need adjustment. The main function of R25,26 is to define a suitable quiescent collector current for the drivers TR6,8, and to pull charge carriers from the output device base when that half of the circuit is in the process of turning off. The value is not normally critical, and may be in the region 47 - 100 Ohms, though I have only tested the circuit for the range 68 - 100. Lower values give faster output device turn-off, possibly at some expense in basic linearity; this is of course only relevant to the Class-B mode. Several rude things have been said about my output network (ie the Zobel network C6,R18, and the output inductor L1 with its damping resistor R19) but I have left it unaltered as while it may be sub-optimal in some respects, IT WORKS, conferring what appears to be complete immunity to instability with capacitative loading. I suspect that L1 could be reduced in value and size, without any ill-effects, but testing this against all possible real-life loads is very lengthy, so there is a powerful incentive to stick with the output network that worked for your Grandfather... 4 THE SOAR PROTECTION. SOAR (Safe Operating ARea) protection is given by the networks around TR18,TR19. This is a conventional single-slope SOAR system, which effectively draws a straight line protection locus across the Vce-Ic graph. This is much more effective than simple current- limiting, but does not follow well the shape of the SOAR area, which is bounded by two curves, one set by maximum power, and the other by second-breakdown. Fortunately, in this application, with its low rail voltages, maximum utilisation of the transistor SOAR capabilities is not really an issue; the most important thing is to observe maximum junction temperatures in the A/AB mode. The extra complexity of dual-slope SOAR limiting was therefore considered unnecessary. In the positive half, TR18 monitors the current through TR7 (via R22) and also the voltage across it. (via R21) D7,8 prevent spurious conduction of TR18,19 when they are reverse biased collector-emitter by large output voltage excursions. In the negative half of the output stage, TR19 conducts when the protection locus is exceeded. TR17 is a current limiter for the VAS transistor TR4; when protection transistor TR19 conducts large currents can potentially be drawn through TR4. TR17 monitors the VAS current through R38, and turns on to shunt base drive away from the VAS when required. This protection is not required for positive SOAR limiting as TR5 is a current-source with inherently controlled output current. There is actually a subtle trap waiting for those who apply SOAR limiting incautiously; TR18,19 can conduct just a trifle in normal operation, and this is usually not very symmetrical so it looks like second-harmonic distortion. You can drive yourself mad looking for the source of this distortion in the VAS or the output stage, and so I strongly recommend that D7,D8 are omitted until the circuit linearity has been checked. 5 GENERAL. It is often stated in hifi magazines that both valve and semiconductor amplifiers sound better after hours, days, or possibly even months, of warm-up time. However, regular readers will have concluded by now, much as I have, that if an unsupported statement appears repetitively in hifi magazines it is almost certainly not true, and I fear this is just another example. It would certainly be possible to come up with a transistor amplifier with bad crossover distortion and a very poorly-controlled quiescent current, which might produce audible changes over time, but the effect would need to be so gross that it would be impossible to miss in even the most casual measurements, and I have never seen this reported as an explanation. Such a case would have to represent plain incompetence in design, for there are a wealth of electronic techniques for rendering the circuit performance more or less immune to variations in beta and temperature. Since this sort of accusation is usually applied with particular force to solid-state Class-A designs, because it is clear that the large heatsinks required take some time to reach their final temperature, I think it important to state emphatically that in this design the Class-A electrical operating conditions stabilise in less than a second, giving the full intended performance. No "warm-up time" beyond this is required; obviously the heatsinks may take a long time to reach thermal equilibrium with ambient, but, as described in the notes above, stern measures have been taken to ensure that device/heatsink temperature has no effect on operating conditions. DC PROTECTION. The published design is not intended as a purely cookbook project- such are not the domain of Electronics World- and therefore omits the important ancillary of relay muting and DC offset protection under fault conditions. It is VERY strongly recommended that a DC protection system is added; eg a ready- designed version such as the Maplin version, which is known as Velleman Kit K4700 (Order Code VE24B) If you intend to design your own relay protection system- and this is fairly straightforward- then here are some points to ponder. Having paid for the relay, for DC protection, it seems sensible to use it for system muting as well, to prevent thuds and bangs from the upstream parts of the audio system from reaching the speakers at power-up and power down. The amplifier itself, being dual-rail (ie DC-coupled) does not generate large thumps itself, but it cannot be guaranteed to be completely silent. Your relay-control system should: 1 Leave the relay de-energised when muted. At power-up, there should be a delay of at least 1 second before the relay closes. This can be increased if required. 2 Drop out the relay as fast as is possible at power-down, to stop the dying moans of the preamp etc from reaching the outside world. My own preference is to do this by sensing the incoming AC; when this disappears, the relay is dropped out within 10 msec, which should be long before the various reservoir capacitors in the system can begin to discharge. However, if the mains switch contacts are generating RF that is in turn reproduced as a click by the preamp, then this system may not be fast enough to mute it. 3 Drop out the relay as fast as is possible when a DC offset of more than 1 - 2 V, in either direction, is detected at the output of either power amp channel; the exact threshold is not critical. This is normally done by low-pass filtering the output (47K and 47 uF works OK) and applying it to some sort of absolute-value circuit to detect offsets in either direction. The resulting signal is then OR-ed in some way with the muting signal mentioned above. 4 Don't forget that the contacts of a relay have a much lower current rating for breaking DC rather than AC. This is an issue that doesn't seem to have attracted the attention it deserves. Please note that the HT rail fuses are intended only to minimise amplifier damage in the event of output device failure. They must not be relied upon for speaker protection against DC offset faults. Fuses in series with the output line are sometimes recommended for DC offset protection. It appears to be true that they have a better chance of saving expensive loudspeakers than the HT fuses, but there are at least three snags: 1 Selection of the correct fuse size is not at all easy. If the fuse rating is small and fast enough to provide some real protection, then it is likely to be liable to nuisance blowing on large bass transients. A good visual warning is given by behaviour of the fuse wire; if this can be seen sagging on transients, then it is going to fail sooner rather than later. At least one writer on Class-A amplifiers gave up on the problem, and coolly left the tricky business of fuse selection to the constructor! 2 It has been widely reported that fuses running within sight of their rated current generate distortion at LF due to changes in resistance caused by RMS heating; this is presumably third harmonic. It is of course true that many widely reported things in the audio field have no more existence than the unicorn, but the mechanism does (for once) sound plausible, so although I have no data on this to offer myself, I think it desirable to make the point. This problem can, in theory at least, be sidestepped by putting the fuse inside the global feedback loop; however what will the amplifier do when its global feedback is abruptly removed when the fuse blows? 3 Fuses obviously have significant resistance (otherwise they wouldn't blow) so putting one in series with the output will degrade the theoretical damping factor. However, whether this is of any audible significance is very doubtful. There is naturally a great deal more that could be said about amplifier protection, and if I am spared I hope to deal with some of the lesser-known issues in a future article. POWER SUPPLY ISSUES. The amplifier design as published yields 20W rms into 8 Ohms, but may be configured for other powers by appropriate choice of supply rail voltage and quiescent current. I gave an upper limit of 30W in the first part of the article, but on mature reflection this seems a bit high for one output pair, and it might be best to reduce the quiescent current somewhat from the 6-Ohm case. The choice of the appropriate quiescent current for a given output voltage capability depends on an appreciation of the load impedances that are likely to be encountered in real life; the published design assumes that a current capability that allows 6- Ohm resistive loads to be fully driven is adequate. Remember that unlike some Class-A amplifiers, this one does not run abruptly into horribly audible current-clipping when it runs out of push-pull Iq into a low-Z load, but goes instead into a low-distortion Class-AB. In general, if the supply rail voltages are increased by, say 10%, then the Class-A/AB mode quiescent current must be similarly increased by 10% to maintain the same load-driving capability. Note that the quiescent dissipation in this mode has now increased by approx 20%, as we have upped both voltage and current, and this needs to be taken into account when arranging the heatsinking. This design has excellent supply-rail rejection, and so a simple unregulated supply is perfectly adequate. The use of regulated supplies is definitely unnecessary, and I would recommend strongly against their use. At best, you have doubled the amount of high- power circuitry to be bought, built, and tested. At worst, you could have intractable HF stability problems, peculiar slew- limiting, and some expensive device failures. Just say no! If you wish to modify the unregulated PSU design given, then remember that in the Class-A/AB mode, a heavy current is drawn continuously; rectifier power dissipation and reservoir capacitor ripple-current capability must be taken much more seriously than is usual for Class-B. In general, 10,000uF is an absolute minimum for the reservoirs, and 20,00uF is strongly recommended. Increases beyond this will do no harm (so long as the turn-on surge can be handled by the rectifiers) and will marginally increase the unclipped output power. When selecting the value of the mains fuse in the transformer primary circuit, remember that toroidal transformers take a large current surge at switch-on. The fuse will definitely need to be of the slow-blow type. COMPONENTS. As before, we have attempted to configure the PCB to use easily obtainable components, in particular the following, which are all available from Maplin Electronics. Maplin order codes are given for reference: 1) Driver heatsinks. The PCB has mounting holes suitable for heatsink Type-SW38-2 (Order Code JW28F) 2) Fuseholder clips. 20mm Fuse Clip Type 1, (Order Code WH49D) 3) Quiescent-adjust preset. Cermet preset 1K, (Order Code WR40T) 4) Wirewound resistors. 3W "WW Min" (Order Code W+value) 5) Non-electrolytic capacitors; Polyester. (eg Order Code WW41U) 6) Output inductor; 18 swg enamelled copper wire.(Order code BL25C) MECHANICAL LAYOUT & DESIGN CONSIDERATIONS. The mechanical design adopted depends very much on personal taste and resources, but I will offer a few points that need to be taken into account: 1 A Class-A amplifier requires extensive heatsinking with a free convective air flow, and this points toward putting the sinks on the side of the amplifier; the front will carry at least the mains switch and power indicator light, while the back carries the in/out and mains connectors, so only the sides are completely free. The internal space in the enclosure will require some ventilation to prevent heat buildup; slots or small holes are desirable to keep foreign bodies out. Avoid openings on the top surface as these will allow the entry of spilled liquids, and increase dust entry. BS415 is a good starting point for this sort of safety consideration, and this specifies that slots should be not more than 3mm wide. 2 A toroidal transformer is strongly recommended because of its low external field. It must be mounted so that it can be rotated to minimise the effect of what stray fields it does emit. Most suitable toroids have single-strand secondary lead-outs, which are too stiff to allow rotation; these can be cut short and connected to suitably-large flexible wire such as 32/02, with carefully sleeved and insulated joints. One of our prototypes had an ILP toroid mounted immediately adjacent to the TO3 end of the amplifier PCB; however complete cancellation of magnetic hum (output level below -90 dBu) was possible on rotation of the transformer. A more difficult problem is magnetic radiation caused by the reservoir charging pulses (as opposed to the ordinary magnetisation of the core, which would be essentially the same if the load current was sinusoidal) which can be picked up by either the output connections or cabling to the power transistors if these are mounted off-board. For this reason the transformer should be kept physically as far away as possible from even the high-current section of the amplifier PCB. As usual with these transformers, make certain that the bolt through the middle cannot form a shorted turn by contacting the chassis in two places. WIRING LAYOUT AND SEMICONDUCTOR INSTALLATION. The distortion performance of a Class-B or AB power amplifier depends almost as much on the topology and layout of the power and ground wiring as on the subtleties of the circuit design. (Things are much simpler if you can assume that the amplifier will never leave Class-A, but with realistic quiescent currents this is not a safe assumption) This has been taken into account in the PCB layout, but the external wiring has to be the responsibility of the constructor. We therefore give a recommended wiring scheme that has been approved by the designer. (The assumption is made that a simple unregulated supply is used; as noted above, a regulated supply is quite unnecessary and may cause unforeseen complications) 1 There are several important points about the wiring for any power amplifier; see the attached wiring diagram: a: Keep the + and - supply wires to the amplifiers close together. This minimises the generation of distorted magnetic fields which may otherwise couple into the signal wiring and degrade linearity. Sometimes it seems more effective to include the 0V line in this cable run; if so it should be tightly braided to keep the wires in close proximity. For the same reason, if the power transistors are mounted off the PCB, the cabling to each device should be configured to minimise loop formation. b: The rectifier connections should go direct to the reservoir capacitor terminals, and then away again to the amplifiers. Common impedance in these connections superimposes charging pulses on the rail ripple waveform, which may degrade amplifier PSRR. c: Do not use the connection between the two reservoir capacitors as any form of star point. It carries heavy capacitor-charging pulses that generate a significant voltage drop even if thick wire is used. As the drawing shows, the "star-point" is tee-ed off from this connection. This is a star-point only insofar as the amplifier ground connections split off from here, so do not connect the input grounds to it, as distortion performance will suffer. 2 Driver transistor installation. These should be mounted onto their separate heatsinks with silicone thermal washers, to ensure good thermal contact. Use the spring clips intended to hold the package firmly against the sink. Electrical isolation between device and heatsink is not essential, as the PCB makes no connection to the heatsink fixing pads, but you will get it anyway unless the washer is damaged. 3 TO3 power transistor installation. The PCB layout allows the TO3s to be mounted on an aluminium thermal-coupling flange which is bolted to the PCB, and the TO3 pins then soldered directly in. Alternatively, the TO3s can be mounted off-PCB (eg if you already have a large heatsink with TO3 drillings) with wires taken from the TO3 pads on the PCB to the remote devices. These wires should be fastened together (two bunches of three is fine) to prevent loop formation; see above. I cannot give a maximum safe length for such cabling; certainly 8 inches causes no stability problems. The emitter and collector wires should be substantial, such as 32/02, but the base connections can be as thin as 7/02 without problems. It is recommended that the flange is drilled with suitable holes to allow bolts to pass through the TO3 fixing holes, through the flange, and then be secured by nuts and crinkly washers which will ensure good contact with the PCB mounting pads. Insulating sleeves are essential around these bolts where they pass through the flange; nylon is a good material for these as it has a good high- temperature capability. Depending on the size of the holes drilled for the two TO3 package pins, (and this should be as small as practicable to maximise the area for heat transfer) these are also likely to require insulation; silicone rubber sleeving carefully cut to length is very suitable. An insulating thermal washer must be used between TO3 and flange; these tend to be delicate and the bolts must not be over- tightened. If you have a torque-wrench, then 10 Newton/metre is an appropriate upper limit for M3.5 fixing bolts. Do not solder the two transistor pins to the PCB until the TO3 is firmly and correctly mounted, fully bolted down, and checked for electrical isolation from the heatsink. Soldering these pins and then tightening the fixing bolts is likely to force the pads from the PCB. If this should happen then it is quite in order to repair the relevant track or pad with a small length of stranded wire to the pin; 7/02 size is suitable for a very short run. 4 Bias-generator transistor TR13 mounting. Our previous design (The Class-B amplifier; PCB-001) used a double emitter-follower or EF output stage; for this the optimal place to mount sensor TR13 for effective thermal compensation was the top of the TO3 cans, to get as close as possible to the output device junction temperature. This is mechanically awkward, and not necessary here. The TRIMODAL design uses a CFP output stage rather than EF, to increase both output efficiency and linearity. The output device junction temperature is now almost irrelevant, being servo-ed out by the local CFP feedback loop, and in Class-B mode the temperature-sensor TR13 must now aspire to reach the temperature of the drivers instead, which is mechanically much simpler. A position for mounting TR13 on the other side of the TR6 heatsink (HS1) is provided on the PCB, and a second thermal washer and spring clip are required for mounting. If this method of mounting is used, then obviously there is a thermal delay and attenuation between the driver and the sensor, due to the thermal mass and convective losses of the heatsink coming between the two devices. A better solution for optimal thermal compensation is to mount the sensor on top of the driver transistor package, ie on the same side of the heatsink. Flying leads are then run back to the original TR13 position. The standard spring clip has enough give (just) to allow the extra transistor and extra thermal washer to be slipped between it and the driver package. The implications of this improvement will hopefully be further explored in a future article. In Class-A/AB mode the quiescent current is controlled by another negative-feedback loop, which can measure the current it is controlling directly, and so this sort of temperature compensation is not an issue. NOTE: Make sure TR13 is properly in contact with the surface it is sensing. Without thermal compensation the quiescent stability in Class-B will be seriously degraded, though almost certainly not to the point where thermal runaway is possible. SAFETY. The amplifier design presented here is inherently safe in that all the DC voltages are too low to present any kind of electric- shock hazard. However, there are a few points I think the constructor should consider. 1 The supply rails are low-voltage, but the reservoir capacitors store a significant amount of energy. If they were to be shorted out by a metal finger-ring then a nasty burn is likely. If your bodily adornment tends toward the metallic then it should be removed before diving into the amplifier. 2 Any amplifier containing a mains power supply is potentially lethal. The amplifier is unusual and slightly more complex than some, so the risks involved in working for some time on the powered-up chassis must be considered. The metal chassis MUST be securely earthed to prevent it becoming live if a mains connection falls off, but this presents the snag that if one of your hands touches live, there is a good chance that the other is touching chassis ground, so your well-insulated training shoes will not save you. All mains connections (neutral as well as live) must therefore be properly insulated so they cannot be accidentally touched by finger or screwdriver. My own preference is for double insulation; for example, the mains inlet connector not only has its terminals sleeved, but there is also an overall plastic boot fitted over the rear of the connector, and secured with a ty-wrap. Note that this is a more severe requirement than BS415 which only requires that mains should be inaccessible until you remove the cover.(With a tool, though a coin is permissible) 3 A Class-A amplifier runs HOT and the heatsinks may well rise above 70 degC. This is not likely to cause serious burns, but it is painful to touch. You might consider this point when arranging the mechanical design. 4 Note the comments on slots and louvres in the section on "Mechanical Design" above. 5 Readers of hifi magazines are frequently advised to leave amplifiers permanently powered for optimal performance. Unless your equipment is afflicted with truly doubtful control over its own internal workings, this is quite unnecessary. While there should be no real safety risk in leaving a soundly-constructed power amplifier powered permanently, I see no point and some risk in leaving unattended equipment powered; in Class-A/AB mode, there may of course be an impact on your electricity bill... TESTING AND FAULT-FINDING. 1 By far the most important step to successful operation is a careful visual inspection before switch-on. As in all power amplifier designs, a wrongly-installed component may easily cause the immediate failure of several others, making fault-finding difficult, and the whole experience generally less than satisfactory. It is therefore most advisable to meticulously check: That the supply and ground wiring is correct. That all transistors are installed in the correct positions. That the drivers and TO3 output devices are not shorted to their respective heatsinks through faulty insulating washers. That the circuitry around the bias generator TR13 in particular is correctly built. An error here that leaves TR13 turned off will cause large currents to flow through the output devices and may damage them before the rail fuses can act. I recommend that the initial testing is done in Class-B mode. There is the minimum amount of circuitry to debug (The Class-A current-controller can be left disconnected, or not built at all until later) and at the same time the Class-B bias generator can be checked for its operation as a safety-circuit on Class-A/AB mode. 2 The second stage is to obtain a good sinewave output with no load connected. A fault may cause the output to sit hard up against either rail; this should not in itself cause any damage to components. Since a power amp consists of one big feedback loop, localising a problem can be difficult. The best approach is to take a copy of the circuit diagram and mark on it the DC voltage present at every major point. It should then be straightforward to find the place where two voltages fail to agree; eg a transistor installed backwards usually turns fully on, so the feedback loop will try to correct the output voltage by removing all drive from the base. The clash between "full-on" and "no base-drive" signals the error. When checking voltages in circuit, bear in mind that C2 is protected against reverse voltage in both directions by diodes which will conduct if the amplifier saturates in either direction. This DC-based approach can fail if the amplifier is subject to high-frequency oscillation, as this tends to cause apparently anomalous DC voltages. In this situation the use of an oscilloscope is really essential. An expensive oscilloscope is not necessary; a digital scope is at a disadvantage here, because HF oscillation is likely to be aliased into nonsense and be hard to interpret. 3 The third step is to obtain a good sinewave into a suitable high-wattage load resistor. It is possible for faults to become evident under load that are not shown up in Step 2 above. Setting the quiescent current for any Class-B amplifier can only be done accurately by using a distortion analyser. If you do not have access to one, the best compromise is to set the quiescent voltage-drop across both emitter resistors (R16,17) to 10mV when the amplifier is at working temperature; disconnect the output load to prevent DC offsets causing misleading current flow. This should be close to the correct value, and the inherent distortion of this design is so low that minor deviations are not likely to be very significant. This implies a quiescent current of approx 50 mA. 4 When you are satisfied that the amplifier is working correctly in Class-B, the Class-A/AB mode can be tested. The voltage across both output emitter resistors R16,17 should now be increased to 300mV (the same as the voltage across R31) and it should remain very nearly constant as the amplifier warms up. 5 It may simplify faultfinding if D7,D8 are not installed until the basic amplifier is working correctly, as errors in the SOAR protection cannot then confuse the issue. This demands some care in testing, as there is no short-circuit protection. INSTALLATION IN CHASSIS. Two mounting holes for fixing the PCB to the chassis are provided on the input edge. These accept standard plastic pillars. Four further fixing holes are provided for fixing the PCB to the heatsink; if the power transistors are mounted off-board then the outer two of these can be used for two more mounting pillars. PLEASE NOTE. Since the component selection, construction, and usage of this PCB are entirely outside our control, we can accept no responsibility for the functioning or performance of amplifiers constructed with it. We are therefore unable to enter into correspondence regarding faultfinding, substitute components, etc. We can accept no liability for loss, damage or injury incurred by the construction or operation of this design.
ADDENDUM TO TRIMODAL AMPLIFIER NOTES. 19 JUNE 95 PCB002A1.DOC Some minor corrections and the fruits of experience... 1) Final testing showed that R22 was not required. It was therefore omitted from the published circuit, though there is still a position for it on the PCB. Fit a wire link in this position. 2) Owing to a production oversight, there are three pads (on links LK9,10) which are covered in solder resist. These should be scraped clean with a sharp knife. Do it just before soldering so the copper does not have time to oxidise. 3) If high-beta input transistors with the ECB pinout are used, it will be necessary to cross the C and B legs before fitting them in the PCB. The obvious insulation to use is 1mm bore silicone sleeving- but actually this is not a good idea. The square-section transistor legs seem to be able to cut through this with surprising ease, and I have found there is every chance of installing two short-circuits along with the components. Two simultaneous faults make faultfinding difficult, so be warned... 4) A Test Mode jumper position (J1) has been added next to the preset; this has the correct pitch for standard pins and push-on jumpers. The jumper must be removed once you are happy the Class-A current controller is working, and the preset turned back to the appropriate position for Class-B biasing. If you don't remove it, TR13 is still active as a Vbe-multiplier in Class-A mode, and it will be impossible for enough quiescent current to flow. 5) There is a minor error in Part 2 of the article. On the second page the use of a 2.56V band-gap reference is discussed; please note that R30 must be increased to 7K (not 5K as stated) in order to get the desired 300mV across R31. 6) Detailed evaluation of the PCB has shown that the grounding of C12 could be improved. If the track to the +ve terminal is cut, and a wire run back from here directly to the 0V supply pad, the -ve HT ripple rejection is improved by several dB. Not essential, but recommended for the perfectionist. 7) As with all Class B or AB amplifiers, layout of supply wiring is critical if the best distortion performance is to be obtained. In PCB tests, the best results were got by taking the +ve HT wire directly off to the supply, while the -ve HT wire was run up the PCB parallel with R16,17 (NB these get hot) and fastened it to the +ve wire at the top PCB edge. The 0V cabling run appeared to be much less critical, and was put into the same fastening; the 3 wires then go to the PSU reservoirs in a tight bundle. Douglas Self.